1. Field of the Invention
The present invention relates to the field of power converters of switched-mode type. Such converters use an inductive element, associated with a power switch and with a free wheel diode, to perform a power conversion and a correction of the power factor, generally based on a D.C. input voltage. Voltage step-down converters (BUCK), voltage step-up converters (BOOST), and buck-boost converters are known.
The present invention more specifically relates to a circuit for helping the switching of the power switch of a switched-mode converter.
2. Discussion of the Related Art
FIG. 1 shows the simplified diagram of a conventional step-up converter 1. Such a converter includes an inductance L0 in series with a free wheel diode DL between two positive input and output terminals 2 and 3 of the converter, the cathode of diode DL being connected to terminal 3. A power switch K connects the midpoint 4 of this series connection to a terminal 5 of application of a negative or reference voltage (generally, the ground) common to the converter input and output. A D.C. supply voltage source 6 provides a voltage VE across terminals 2 and 5. On the output side, a storage capacitor C0 generally connects terminals 3 and 5 and provides a voltage VS to a load Q. Load Q has been shown in FIG. 1 by dotted lines integrating capacitor C0, which may or not belong to the load. Switch K is controlled by a circuit 7 (CTRL), for example, in pulse-width modulation (PWM).
The operation of a step-up converter will now be described. When switch K is on, power is stored in inductance L0 and load Q is supplied by the power stored in capacitor C0. When switch K is off, inductance L0 gives back the stored power to capacitor C0 via free wheel diode DL.
FIG. 2 shows the simplified electric diagram of a step-down converter 1′. It shows the same components as in FIG. 1. However, here, switch K is connected in series with inductance L0 between positive input and output terminals 2 and 3. Free wheel diode DL grounds the junction point 4′ of switch K and inductance L0, its cathode being connected to point 4′. Switch K may also be provided between the negative terminal of source 6 and the anode of diode DL.
The operating principle is the same. Power is stored in inductance L0 during the on periods of switch K. During periods when switch K is off, this power is given back to capacitor C0, free wheel diode DL being used to loop back the circuit.
A problem which arises with switched-mode converters, also called hard-switching converters, in which the current and the voltage cross each other upon each switching, is linked to the switch turning-on.
Indeed, upon each turning-on of switch K, free wheel diode DL must block. Now, at the blocking of a diode, especially of a PN junction diode, a recovered charge phenomenon occurs.
This phenomenon is illustrated by FIGS. 3A to 3C, which show, in relation with the circuit of FIG. 1, an example of the shape of current IDL in the free wheel diode, of output voltage VS and of current IT in switch K.
Switch K is initially assumed to be off. Accordingly, a current ILf flows through diode DL. This current corresponds to the power given back by inductance L0. The output voltage is at a level V0. As for switch K, the current IT flowing therethrough is null.
It is assumed that at a time t1, control circuit 7 turns switch K on. During the switching, current IL in the inductance, which corresponds to the sum of currents IDL and IT is a constant. Accordingly, the current which, during the switching, increases in the switch, translates as a decrease with an inverse slope of the current in diode DL.
At a time t2, the current in diode DL becomes zero and the current in the switch reaches level ILf. At this time starts the recovered charge phenomenon of diode DL. This known phenomenon translates as an inversion of the current through the diode to reach a level IRM corresponding to the maximum recovery current of the diode. Current IRM is reached at a time t3 from which the current through the diode tends towards zero again, reaching it at a time t4. Since the current in inductance L0 is, during the switching, substantially constant, the negative current peak on the diode side translates as an overcurrent in switch K, the maximum value of which corresponds to current ILf plus value IRM. On the side of voltage VS, the voltage decrease in practice intervenes from time t3, that is, from the inversion of the current slope in diode DL. In other words, the voltage across the diode is zero between times t2 and t3 corresponding to the first recovery phase ta. It can be considered that the diode then transiently conducts in reverse. Between times t3 and t4 (second recovery phase tb), voltage VS decreases from V0 to a zero voltage. The voltage provided to capacitor C0 is here considered. Indeed, the presence of the capacitor in practice results in output voltage VS remaining approximately stable.
The slope between times t1 and t3 of the current decrease in diode DL depends on the turn-on speed of the switch and thus on its di/dt at the turning-on. The higher this di/dt, which favors an abrupt switching, the higher amplitude IRM is for a PN-junction diode. However, the smaller di/dt, the longer the recovery time at the blocking (trr=t4−t2).
The losses in a diode according to the di/dt value have a parabolic shape. There is an optimal point where the surface area of the current shape between times t2 and t4 is minimum, which results in minimum losses of recovered charges in the diode.
For switch K, the recovered charge phenomenon of the diode is particularly disturbing. Indeed, for a step-up converter, the switch then sees across its terminals, between times t2 and t3, output voltage VS. In the case of a step-down converter, the voltage seen by the switch across its terminals corresponds to the voltage of generator 6. In all cases, it is the highest voltage between voltages VE and VS.
High losses can then be observed in switch K. In FIGS. 3A to 3C, the loss periods have been symbolized by hatching on the various timing diagrams.
In practice, the losses in switch K (generally, a power transistor) at its turning-on (times t1 to t4) form most of the switching losses of the converter. In particular, the losses due to the actual blocking of the diode and the turn-off losses of the switch are negligible with respect to the losses generated at its turning-on.
A first solution to reduce this disadvantage consists of using diodes with no recovered charges, for example, Schottky or SIC-type diodes.
A first disadvantage of this solution is that diodes with no recovered charges are often limited to a breakdown voltage of some hundred volts. This solution is thus not applicable to converters operating under voltages of several hundreds of volts, which is in practice current in power electronics. Several diodes in series must then be provided to increase the breakdown voltage.
Another disadvantage of this solution is that, even if it decreases losses linked to recovered charges (times t2 to t4), the most significant losses linked to the sole switch turning-on are not avoided. Referring to the example of FIGS. 3A to 3C, the use of a diode with no recovered charges results in an zero voltage VS from time t2. There thus remain the losses linked to the surface areas located between times t1 and t2.
Another disadvantage of diodes with no recovered charges is that they are particularly expensive as compared to PN diodes. Presently, the cost ratio is greater than 20.
A second solution to attempt solving recovered charge problems is to provide a circuit for helping the switching of the power switch of the converter.
FIG. 4 shows a conventional example of such an aid circuit, applied to a step-up converter such as shown in FIG. 1. FIG. 4 shows all elements of FIG. 1, to which is added a circuit 8 for helping the switching of switch K. This circuit is formed of an inductance L, associated in parallel with a resistor R and a diode D, between point 4 and switch K. The function of inductance L is to control the switch di/dt. By decreasing this di/dt value, amplitude IRM is decreased.
A problem which arises is that resistor R must be provided to dissipate a reverse overvoltage in inductance L. Indeed, upon the turn-on switching of switch K, the voltage across inductance L takes the value of output voltage VS. The same losses occur at the transistor turning-off. These are resistive losses which are all the greater as the di/dt value is high. In other conventional examples, dissipation element R is replaced with a capacitor, a zener diode, etc.
Thus, this second solution has the same disadvantages as the use of a diode with no recovered charges.
A third known solution (not shown) consists of a circuit for helping the switching using the transient switching resonance. Such a circuit uses, like the circuit of FIG. 4, an additional inductance. However, to avoid resistive loss problems, a second switch, the control of which is desynchronized with respect to that of switch K, is used.
An example of a switching aid circuit of this type is described in paper “An overview of soft switching technics for PWM convertors” by G. Hua and F. Lee, published in EPE Journal, Vol. 3, March 1993.
Such a solution provides satisfactory results, but has a particularly complex and expensive implementation. In particular, a control system desynchronized from the used switches must be provided. Further, as compared to the circuit of FIG. 4, it is necessary to have an additional power switch, two additional diodes and, above all, a high-voltage capacitor.
The present invention aims at overcoming the disadvantages of known switching aid circuits.